Even with finest modern materials, the output transformer remains the weakest link. At low frequencies its finite inductance steals current away from the load, and at high frequencies its stray capacitance, leakage inductance, and high-frequency copper losses combine to attenuate amplitude and introduce severe phase shifts. From the early days of hi-fi, audio designers strove to create the ideal amplifier. Much effort went into optimizing power transformer design, and even now, the seminal text books on transformers and basic magnetic analysis date back to this early time.
H.S Black discovered negative feedback for electronics in 1934, and it was immediately realized that feedback could go a long way to curing many of the problems introduced by the output transformer and other the electronic amplifying devices. In fact, it was felt that the quality of an amplifier was limited only by the amount of negative feedback that could be applied.
How much feedback that can be realistically appied is limited by how much phase shift exists from input to output of a system. Most of the feedback-limiting phase shift in a vacuum tube amplifier occurs in the output transformer; therefore, if only it could be eliminated, large amounts of negative feedback could be employed, thus creating the perfect amplifier. Today we realize that negative feedback is not the panacea it was once thought to be. In small to moderate doses it can be beneficial, but, as solid-state amplifier designers eventually discovered a generation later, too much feedback always sounds less musical than too little feedback.
As with any problem in engineering, the optimal solution is not to fix it with additional circuitry, but to remove it altogether. It seems logical then, that if an amplifier can be no better than its output transformer, eliminating the transformer altogether should result in superior performance. Hence the quest for the holy grail of tube amplifier design that has fascinated and tantalized audiophiles for over 40 years: how to design a rugged and powerful Output TransformerLess (OTL) amplifier.
Plate impedance alone does not completely determine the available power. It is also affected by whether your choice of circuit forces you into Class A operation (all power tubes always conduct), or whether Class AB (half the tubes may cut off during a half cycle) or Class AB2 (same as AB, but grid is driven positive, resulting in grid current flow) are also possible. However, once the class of operation is determined, available power is determined by plate resistance.
Even using several of the best tubes available, plate resistances are still considerably higher than load resistances. That's OK, you can build an OTL amplifier; it will just be relatively inefficient. For example,
a single 20 Ohm plate resistance tube supplying a 4 ohm
load can never be better than 16% efficient (=4/(20+4) x 100%).
Reasonable power can still be provided to the 4 ohm load,
but it requires a higher supply voltage than the peak output voltage,
and you have to accept this and design for the dissipation of the circuit.
While we cannot change the inherent plate characteristics of a tube, we can change the way the tube is arranged and driven in an amplifier circuit. This can have a dramatic effect upon the output impedance of the amplifier which tells us how stiff a voltage source we have made. This is also expressed by some as the damping factor of an amplifier - the lower the output impedance, the better voltage source you have achieved. The amplifier will then be better able to handle the nonlinearities, distortions, and back emf produced by the usually very complex speaker load.
The common available topologies for OTL amplifiers have some very important
differences in the output impedance characteristics which must be carefully
considered when selecting which topology you want to use. Although the power
capabilities of the different topologies may remain unchanged when they operate in the
same class (A, AB or AB2), the sonics may be substantially different due to the wide
range of output impedance that can be achieved.
For the sake of this paper, the 6C33C-B tube will be used for numerical example. At a bias voltage of 145 V, and a plate current of 400 mA, this tube has a plate resistance of 100 ohm, and a gain of 2.7.
At lower currents, the plate resistance increases, a factor to bear in mind when selecting the bias point of the tube. At higher currents, the plate resistance drops - fortunately so, otherwise even the 6C33C-B would be horribly inefficient in supplying a 4 ohm load, and a high-power OTL amplifier would not be practical. At a plate voltage of 70 V, and 1 A current, the resistance of this tube drops to 40 ohm.
This changing plate impedance is another reason to strive to get the output impedance
level as low as possible. If the
output impedance is high relative to the load, and it changes substantially with current,
a significant amount of
odd harmonic distortion is created.
Fig. 1: Cathode Follower OTL
In order to prevent dc current from flowing through the speaker load of this configuration, the speaker can be paralleled with a large inductor which will shunt the dc bias current. This is still an OTL - there is no transformer between the power tube and the load, and none of the detrimental effects of the transfer function from input to output of a transformer coupled load. The most important benefits of OTLs are eliminating the transformer wire, and connecting the speaker directly to the power tube with no other elements in series. However, the low-frequency loading effect of the inductor will detract from the bass performance of this configuration.
This amplifier must, of course, operate in Class A regime only.
Since we are concerned here with much higher output
power designs, this topology will not be considered further.
Fig. 2: Common Cathode OTL
As with the cathode follower OTL, the problem of dc current in the load can be mitigated by an inductor
in parallel with the load. Also, it must be operated Class A, so it is of little real interest in designing a
high power OTL
Fig. 3: Push-Pull Cathode Follower OTL
Using two tubes in parallel in the cathode follower configuration of the previous section would give Rp(2+2u), the lowest achievable open loop output impedance using a pair of tubes. The push-pull cathode follower output impedance is four times the theoretical minimum because the two cathode followers are in series, not in parallel.
If four 6C33C-B tubes were used in this particular configuration, biased at
145 V and 400 mA each, the open-loop (i.e. before any feedback is considered)
output impedance would be 27 Ohms.
Fig. 4: Single-Ended Push-Pull OTL
In the SEPP output stage the load appears in the cathode circuit of the upper tube and in the plate circuit of the lower tube. If a conventional push-pull ground-referenced driver stage is employed, the upper tubes behave as cathode-followers while the lower tubes act as common-cathode amplifiers.
This creates an imbalance of gain and output impedance between the upper and lower stages, unacceptable in good amplifier design. If true push-pull operation is to be achieved, the upper and lower tubes must be driven with equal but out-of-phase signals, but the presence of the load in the input circuit of the upper tubes makes this difficult to achieve in practice.
In a conventional push-pull amplifier using solid state devices, this problem of balance is overcome
by using a P-channel device. Since we only have single-sex electron tubes,
and positron tubes do not exist (except theoretically, and perhaps amongst
audiophiles of the future on the Starship Enterprise), the SEPP cannot be driven
directly, and it must always be considered in conjunction with a suitable drive scheme.
Fig. 5: Futterman OTL
This arrangement results in equal but opposite drive signals being applied between the grid and plate of both sets of tubes. He went on to claim that when so driven, the output tubes are operating as cathode followers. Unfortunately, this is not correct, and Futterman's driver design did not achieve cathode follower operation. This is because the 100% positive feedback introduced by standing the phase splitter on the output voltage effectively canceled the 100% negative feedback from the load that produces the cathode follower effect in the first place.
In other words, Futterman succeeded in converting the upper tube into a common-cathode amplifier, thus matching the behavior of the lower tube, and in so doing lost the cathode-follower impedance advantage! Instead of getting the desired theoretical output impedance of two parallel cathode followers, Rp/(2+2u), the Futterman OTL has the output impedance of two parallel common cathode amplifier tubes, Rp/2.
Futterman's solution to provide balanced drives to the SEPP is best known in the US. In Japan, OTL designers mostly used a cathode-coupled phase inverter driver first proposed by Hiroshi Ameniya in 1955. While this driver provided lower drive impedance and twice the signal swing of the split-load phase inverter, it too resulted in the output tubes behaving as common-cathode amplifiers with their attendant high source impedance of Rp/2.
It is important to note that the Futterman and other schemes proposed for the SEPP drive use feedback to achieve the balanced signals. But even with a high gain tube driving the output stage, the output impedance is relatively high, so the feedback only serves to balance the drives. In this regard, the gain of the driver tube is not fully utilized to achieve full benefits.
If four 6C33C-B tubes were used in the Futterman OTL, biased at 145 V and 400 mA each, the open-loop output impedance would be 25 Ohms (coincidentally, about the same as the Push-Pull Cathode Follower).
The Futterman circuit, and other solutions to driving the SEPP, allow for Class AB operation. Care must
be taken if pushing into Class AB2 since the driver circuits are not balanced perfectly, and grid
current will introduce assymetry between the top and bottom tubes, and resulting distortion.
Fig. 6: Variation of Futterman OTL
If four 6C33C-B tubes were used in this circuit topology, the open-loop output impedance would be about 6.8 Ohms, 4 times lower than the original Futterman!
Of course, the lower output impedance does NOT mean that the variation on the Futterman can provide four times as much power. The power handling capabilities of both circuits remain the same, since it is the tube plate characteristics that determine power, as discussed earlier. The circuits will need different drive requirements to get there, but they can both run at the same power level.
Also, this low output impedance could only be achieved with a high gain tube. If pure triode operation were desired for the amplifier,
and a realistic gain tube were used, the output impedance would be closer to 3 times lower than the original Futterman circuit.
Fig. 7: Circlotron OTL
With the Circlotron arrangement each output tube now behaves in an identical manner. Half the signal across the load appears in the cathode circuit of each tube resulting in partial cathode-follower operation. While the output impedance of this circuit is somewhat higher than the best optimal cathode follower achieved with the Futterman Variation, Rp/(2+u) versus Rp/(2+2u), it is still considerably better than the Rp/2 with identical tubes in a conventional Futterman. Best of all, a conventional push-pull driver stage will give achieve perfect balance with the circlotron, without resorting to circuit tricks, and no feedback around a high-gain tube is used in the driver stage.
If four 6C33C-B tubes were used in the circlotron topology, the open-loop output impedance would be about 10.6 Ohms. Again, not quite as good as what can be achieved with the Variation on the Futterman, but usable.
There are several other advantages to the Circlotron. Both sets of cathodes are at ground potential and can therefore be biased from a common negative voltage. With the SEPP, two bias voltages are required, one ground referenced and the other referenced to the negative rail.
Also, because of the asymmetrical arrangement of the output tubes with respect to the rail voltages, any ripple, power line or signal-induced transient voltages are fed directly into the grid circuit of the lower tubes, via the cathode, where it gets amplified along with the input signal. The upper tubes are immune from this as the rail voltage looks into the plates and therefore does not modulate the grid-to-cathode voltage. The result is a significant hum component at idle and signal-dependent DC offset at higher signal levels. The only way to prevent this from happening is to provide for electronic regulation of both power supply rails, although hum can be reduced significantly with high negative feedback.
With the Circlotron arrangement, ripple and noise on the rail supplies do not affect grid bias. Any modulation of the rails affects only the tubes' plate voltage. Ripple and power line induced transients are automatically canceled in the load, since the load only responds to differential inputs, and these disturbances are common-mode. This is not true for the Futterman circuits.
Given the benefits of the Circlotron: low source impedance, balanced operation and high common-mode rejection, it is possible to dispense with negative feedback altogether. The Atma-Sphere MA-1 is one product that does this. However, despite using 12 6AS7G's in a Circlotron output configuration, however, the damping factor is still too low for good compatibility with many of the lower impedance speakers available. Realizing this, Atma-Sphere offers a tapped auto-transformer to match the 11 ohm output impedance of the MA-1 to a one, two, three or four ohm load.
For many tube users, the concept of adding a transformer to the
OTL topology defeats the whole purpose of the exercise - Sorry folks, if it
needs a transformer to work properly, it ain't an OTL! It may still sound very
good, and the auto-transformer design may be an improvement over other approaches,
but it's not the holy grail. (Note: it may also be a lot more reliable than a pure OTL -
an important consideration for anyone making these amplifiers for a living.)
Adding some feedback to the circlotron has the added advantage of correcting for the
distortion created in the output stage under large signal excursions. It also
permits low-distortion Class AB2 operation of the output tubes. If the driver
impedance is kept low and can source sufficient current, it is possible to get
almost twice the power that can be achieved with class AB1 operation. But
even with low driver impedance, high open-loop distortion is the result when
positive grid current begins to flow. If left uncorrected it appears as if the output
waveform is clipping. By applying corrective feedback the driving voltage is
automatically increased to compensate for this effect, and the full potential of the
output tubes can be realized.
Until recently the best power triode for OTL's was the 6336A. Very popular in Japan and France, it was even mentioned favorably by Futterman in his famous 1954 paper in the JAES. Bothered by the high grid emission of the 6AS7G, as well as by its low mu of 2, he remarks that the then new Chatham 6336, with its higher mu of 2.7, and its 100 ohm plate resistance (with both sections in parallel) performed well in his design.
The 6336A was exceeded only by the Sylvania 7241 triple-triode in low plate resistance. Rated as having a plate resistance of 67 ohms at 550mA, the 7241 never caught on overseas because it was expensive and hard to find. Another tube with limited availability was the EC33C. Made in the Soviet Union especially for the Japanese market (according to Jean Hiraga in "Les amplificateurs O.T.L.") it was presumably a forerunner of the by now famous Sovtek 6C33C-B.
Previously only available for Soviet military use, the 6C33C-B power triode
is now readily obtainable in the West, ever since the fall of the Iron Curtain.
Very similar in performance to the 6336A but far cheaper, this tube is built like a
tank with its thick glass and internal bracing to reduce microphonics.
It has quickly become the tube of choice for OTL designers around the world.
As used in the Covi Mark II Circlotron OTL, four 6C33C-B tubes generate over 100W into 4 ohms and over 125W into 8 ohms running class AB2 with a plate voltage of 145V. At the sine wave peaks, that's a real 200 W instantaneous, into 4 ohms, which means each 6C33C-B is delivering 3.75 A! Output power can be increased even further (with a corresponding reduction in reliability) with higher plate voltages. For some comments on how reliable this circuit and these tubes are, go to 6C33C-B OTL amplifier reliability.
Further details of this output stage of this amplifier are given on the
OTL Amplifier Output Stage Schematic
The input stage is described in detail on the OTL Amplifier Input Phase Splitter Schematic web page.
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Original: 12-11-96, revised 1-22-97